Transimpedance amplifiers are used to amplify and convert current signals to voltage signals. The input current source typically shows high impedance at low frequencies, which in most applications is shunted by parasitic capacitances. These capacitances lower the source impedance at higher frequencies. As a result it is advantageous for a transimpedance amplifier to provide low input impedance over a wide frequency range.
In broadband fiber-optical data transmission systems, transimpedance amplifiers are, for example, driven by photodiodes. In dependence of the light intensity at the fiber input and dependent on the length and the quality of the fiber interconnect the current generated by the photodiode can vary by several orders of magnitude.
In order to achieve good noise performance, the transimpedance of the amplifier needs to be maximized as far as possible, which is typically limited by the amplifier bandwidth requirements. State of the art transimpedance amplifiers in today's data transmission systems require low supply voltages, e.g. 5V, 3.3V or below. These three requirements—high input-current dynamic-range, high transimpedance and low supply voltage, lead to problems in transimpedance amplifiers of the prior art, which will be discussed in detail with reference to the prior art transimpedance amplifiers shown in FIGS. 1a, 1b and 1c. 
The prior art transimpedance amplifier in FIG. 1a consists of an input transistor Q1, a load resistor RL, an output transistor Q2, a constant current source I1 and a feedback resistor RF. The input current is applied to the base of the common emitter transistor Q1. The collector of this transistor is connected to the positive supply voltage VCC using a load resistor RL. The base of the emitter follower transistor Q2 is connected to the collector of Q1. The collector of Q2 is connected to the supply voltage. The emitter of this transistor is driven by the constant current source I1. A voltage controlled current feedback is provided by means of a feedback resistor, which is connected between the emitter of Q2 and the base of Q1. The output voltage VOUT is available at the emitter of Q2.
Assuming a base emitter voltage drop of about 900 mV, which is a typical value for state of the art bipolar circuit technologies, the voltage at the base node of Q1 is 900 mV referred to ground. With no input current signal IIN and neglecting the finite current gain of Q1 (i.e. assuming IBQ1=0), the output voltage VOUT also equals 900 mV. In a typical fiber optical transmission system operating in the Gbps-region, the input current values show a wide dynamic range from e.g. 10 μA up to 2 mA. In order to provide sufficient sensitivity and gain, the feedback resistor RF should be about 5 kΩ. Consequently, at the maximum input current IIN=2 mA, the voltage drop across the feedback resistor RF would be 10V. The corresponding theoretical output voltage would be −9.1V, which is obviously impossible with a single positive 5.0 or 3.3V power supply.
Any limitation of the dynamic range is undesirable for most applications. Also providing dual voltage sources in order to extend the dynamic range is expensive and not practical. Therefore, in order to overcome this problem other state of the art transimpedance amplifiers use MOSFET devices in adaptive feedback loops. These MOSFET devices are coupled across resistors and thus limiting or clamping the output voltage of the transimpedance amplifier. An example of such kind of transimpedance amplifier is disclosed in U.S. Pat. No. 5,532,471: “Optical Transimpedance Amplifier With High Dynamic Range” by H. Khorramabadi et al. FIG. 1 of the named patent is reproduced herein as FIG. 1b (prior art). Such kind of solutions have several drawbacks. For monolithic integration they require a BiCMOS process with both bipolar and CMOS devices being realized on a single substrate, which is a costly solution. Additionally, this solution requires a low frequency adaptation loop, which may lead to insufficient performance if the level of the input current varies rapidly, as in burst mode applications.
The same disadvantages hold for the approach disclosed in U. S. Pat. No. 6,583,671 B2: “Stable AGC Transimpedance Amplifier With Expanded Dynamic Range” by J. G. Chatwin. Which also uses MOSFET devices in parallel to the transimpedance feedback resistor.
A different approach is used in U.S. Pat. No. 5,708,292: “Method And Apparatus For Providing Limiting Transimpedance Amplification” by W. A. Gross. FIG. 2 of the named patent is reproduced herein as FIG. 1c (prior art). the voltage drop across the feedback resistor RF is limited by diodes or preferably Schottky-type diodes. Compared to the MOSFET solution described above, the advantages of this approach are the simplicity, the inexpensiveness and the suitability for input signals with rapidly changing amplitude levels. However, this solution suffers from several disadvantages: While diodes are always available in bipolar technologies, the preferred Schottky diodes do not exist in modem bipolar technologies, because in today's technologies for connection of the devices polysilicon is used below the metallization. The voltage drops across diodes, however, equals the base-emitter voltage drop of bipolar transistors, leading to operating point problems in the circuit given in figure 1c. Furthermore, the diode voltage drop is highly temperature dependent. Therefore in limiting operation the output voltage of the circuit given in figure 1c decreases strongly with increasing temperature.